Removing close-in interferers through a feedback loop

ABSTRACT

System and method for elimination of close-in interferers through feedback. A preferred embodiment comprises an interferer predictor (for example, interferer predictor  840 ) coupled to a digital output of a direct RF radio receiver (for example, radio receiver  800 ). The interferer predictor predicts the presence of interferers and feeds the information back to a sampling unit (for example, sampling unit  805 ) through a feedback circuit (for example, feedback unit  845 ) through the use of charge sharing. The interferers are then eliminated in the sampling unit. Additionally, the number and placement of zeroes in a filter in the sampling unit is increased and changed through the implementation of arbitrary-coefficient finite impulse response filters.

This application claims the benefit of U.S. Provisional Application No.60/348,902, filed on Oct. 26, 2001, entitled “Direct RF Sampling withRecursive Filtering Method”, which application is hereby incorporatedherein by reference.

TECHNICAL FIELD

The present invention relates generally to a system and method of radiofrequency (RF) direct sampling radios, and more particularly to a systemand method for removing interferers that are in close proximity to adesired signal via the use of feedback and changing fromconstant-coefficient FIR filtering to arbitrary-coefficient FIRfiltering.

BACKGROUND

Generally, discrete-time radio frequency (RF) is a newly emerging fieldin wireless digital communications wherein analog continuous-time RFsignals that are transmitted over-the-air are directly sampled into adiscrete-time sample stream suitable for digital signal processing. Atypical wireless digital communications device would use analog filters,duplexers, mixers, analog-to-digital converters (ADC), etc. to convertthe analog continuous-time RF signals into a digital data stream that issuitable for digital signal processing. Unfortunately, analog circuitcomponents, especially components such as capacitors, inductors,resistors, etc. necessary for the analog filters are difficult tointegrate into an integrated circuit. This is especially true for theprecise values of these components required for use in filters. Ofcourse, it is the desire of the manufacturer to maximize the degree ofintegration for the wireless transceivers. This is because the morehighly integrated a wireless transceiver can become, the lower theproduction costs for the transceiver and the transceiver will typicallyuse less power during operation.

Discrete-time RF involves the direct conversion of the analogcontinuous-time RF signal into discrete-time sample stream through theuse of a direct sampling mixer, without having to undergo anyintermediate analog continuous-time filtering, downconversion, etc. Anexample of a direct RF sampling mixer is one that uses current toperform its sampling. The current-mode direct sampling mixer convertsthe received analog continuous-time RF signal into a current that isthen integrated by a sampling capacitor. The charge on the samplingcapacitor is then periodically read to produce the discrete-time samplestream.

The analog continuous-time RF signal being directly converted into adiscrete-time sample stream may often contain more than a desired signallocated in a frequency band of interest (commonly referred to as asignal of interest). In many circumstances, there are interferers alongwith the signal of interest being sampled by the direct RF samplingmixer. The interferers may be the result of noise sources, such as otherradio frequency devices and communications networks operating in closeproximity with the direct RF sampling mixer, large electrical motors,electrical appliances, etc. The interferers may be located relativelyfar away from the signal, close to the signal (commonly referred tocollectively as out-of-band interferers), or they may actually occur atfrequencies that also carry the signal (commonly referred to as in-bandinterferers).

In the case when the interferers are in-band, active interfererdetection and cancellation may be an only option for removing theinterferers. However, when the interferers are out-of-band, filteringcan be used to eliminate the interferers.

Filtering can be used to eliminate interferers that are out-of-band, andcan take place either in an analog domain or a digital domain. Analogfiltering occurs early on, perhaps as early as immediately after theanalog RF signal is received by an antenna. Digital filtering, on theother hand, can only occur after the discrete-time sample stream createdfrom the analog RF signal has been converted into a digital data stream.This implies that any digital filtering that is to take place, mustoccur later in the signal processing sequence.

A problem that is associated with out-of-band interferers is that, whilethey may have no direct impact on the signal of interest, they may besignificantly larger in magnitude than the signal of interest. If thisis the case, then it is required that certain RF front-end electronics,such as amplifiers, have good linearity. Linearity is required so thatthe presence of the large interferers do not distort the performance ofthe RF front-end electronics in such a way that the electronics do notoperate properly on the relatively smaller signal magnitudes of thesignal of interest. If the out-of-band interferers are eliminated, thenthe linearity of the RF front-end electronics can be relaxed due to areduction of the overall dynamic range of the signal being provided tothe electronics, e.g., only the dynamic range of the signal of interestmust be dealt with by the front-end electronics.

The direct conversion of the analog continuous-time RF signal into adiscrete-time sample stream by the direct RF sampling mixer can includea built-in finite impulse response (FIR) filtering operation. The FIRfiltering comes as a result of an accumulation and decimation ofmultiple samplings of the analog RF signal into a single discrete-timesample by a sampling capacitor. However, the direct RF sampling of theanalog RF signal using fixed current gains and constant capacitive loadsmay result in only FIR filter with constant coefficients. In manyoccasions, it is desired that arbitrary-coefficient filtering beavailable to help eliminate interferers, anti-aliasing, etc. Additionalfiltering can be added, but only with the expense of additionalhardware.

One disadvantage of the prior art is that the use of analog filters toeliminate out-of-band interferers can entail the use of high-orderanalog filters if the out-of-band interferers are close to the signal ofinterest. High-order analog filters can be difficult to implement,especially on an integrated circuit.

A second disadvantage of the use of analog filters to eliminatedout-of-band interferers is that while low-order analog filters can berealized relatively easily, but they are likely to not be able to removethe close-in interferers, therefore, the requirement of good linearityin the RF front-end electronics must be maintained.

A disadvantage of constant-coefficient FIR filtering is that thefiltering may have high sidelobes and an insufficient roll-off rate tohelp eliminate aliasing and/or interferers. Additional filtering can beadded to perform these needed tasks, but constant-coefficient filterswith low cut-off frequencies are difficult to realize in integratedcircuits.

SUMMARY OF THE INVENTION

These and other problems are generally solved or circumvented, andtechnical advantages are generally achieved, by preferred embodiments ofthe present invention which use feedback to eliminate close-ininterferers and provides arbitrary-coefficient FIR filtering without theaddition of a significant amount of hardware.

In accordance with a preferred embodiment of the present invention, amethod for providing filtering in a current-mode sampling mixercomprises 1) providing a received radio frequency (RF) current, 2)coupling the RF current to a first load for a first period of time toaccumulate a charge on the first load, 3) decoupling the RF current fromthe first load, 4) coupling the RF current to a second load for a secondperiod of time to accumulate a charge on the second load, 5) reading theaccumulated charge on the first load, 6) decoupling the RF current fromthe second load, 7) reading the accumulated charge on the second load,and 8) repeating 2–8.

In accordance with another preferred embodiment of the presentinvention, a current-mode sampling mixer comprises an amplifier toproduce a current based on a radio frequency (RF) signal, a plurality ofloads, each load switch-ably coupled to the amplifier, each loadcontaining circuitry to accumulate a charge based on the current, and aplurality of charge read-out circuits, each charge read-out circuitswitch-ably coupled to a load, the charge read-out circuit to permitextraction of the charge accumulated on the load.

In accordance with yet another preferred embodiment of the presentinvention, a method for removing interferers comprises creating adiscrete-time sample stream (DTSS) from an analog continuous-time radiofrequency (RF) signal, converting the DTSS into a digital data stream(DDS), predicting a presence of interferers in the DDS, feeding backinformation about the predicted interferers, and eliminating thepredicted interferers from the DTSS.

In accordance with another preferred embodiment of the presentinvention, a radio frequency (RF) radio receiver comprises a signalinput, a sampling unit coupled to the signal input, the sampling unitcontaining circuitry to convert an analog signal provided by the signalinput into a discrete-time sample stream (DTSS), a signal processingunit coupled to the sampling unit, the signal processing unit containingcircuitry to filter and convert the DTSS into a digital data stream(DDS), an interferer predictor unit coupled to an output of the signalprocess unit, the interferer predictor unit containing circuitry todetect and provide information regarding interferers in the DDS, and afeedback circuit having an input coupled to the interferer predictorunit and an output coupled to the sampling unit, the feedback circuitcontaining circuitry to provide the information produced by theinterferer predictor unit to the sampling unit

An advantage of a preferred embodiment of the present invention is thatclose-in interferers are eliminated from a signal of interest throughthe use of a feedback signal. This permits a relaxation of linearityrequirements for RF front-end electronics. Electronics with lowerlinearity requirements are less expensive and easier to realize inintegrated circuits. This relaxation of the linearity of the RFfront-end electronics can be achieved by linearizing a sampling circuitportion of the direct RF sampling circuit itself.

A further advantage of a preferred embodiment of the present inventionis that the need for a high-order analog filter to remove the close-ininterferers have been eliminated and a low-order analog filter can beused in its place to simply remove interferers that are far away fromthe signal of interest. As noted previously, low-order analog filterscan be realized easier and less expensively in an integrated circuit.

Yet another advantage of a preferred embodiment of the present inventionis that the use of feedback to eliminate close-in interferers is thatthe feedback is dynamic, meaning that if the signal of interest and/orthe interferers move, then the feedback can readily reflect this and theinterferer elimination can adjust. The use of analog filters isgenerally not dynamic (since parameters of analog filters can sometimesbe adjusted, but not without restriction) and is normally set once theanalog filters are designed.

Yet another advantage of a preferred embodiment of the present inventionis that the overall power consumption is reduced by reducing thecomplexity of the analog filters and the linearity of the RF front-endelectronics.

An additional advantage of a preferred embodiment of the presentinvention is that the increase in the length of the FIR filtering comeswith minimal additional hardware, therefore, additional filteringperformance is gained at minimal costs.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the invention that follows may be better understood.Additional features and advantages of the invention will be describedhereinafter which form the subject of the claims of the invention. Itshould be appreciated by those skilled in the art that the conceptionand specific embodiment disclosed may be readily utilized as a basis formodifying or designing other structures or processes for carrying outthe same purposes of the present invention. It should also be realizedby those skilled in the art that such equivalent constructions do notdepart from the spirit and scope of the invention as set forth in theappended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawing, in which:

FIGS. 1 a and 1 b are prior art diagrams of current-mode samplingmixers;

FIG. 2 is a prior art diagram of a current-mode sampling mixer withcyclic charge read-out capability;

FIG. 3 is a plot of frequency responses for various constant-coefficientFIR filters;

FIGS. 4 a–d are diagrams of current-mode sampling mixers with samplingcapacitors of differing capacitances and the ability to implementarbitrary-coefficient filtering, according to a preferred embodiment ofthe present invention;

FIG. 5 is a diagram of a current-mode sampling mixer with an array ofsampling capacitors, capable of providing high-resolution approximationof a continuous capacitance curve, according to a preferred embodimentof the present invention;

FIG. 6 is a diagram of a current-mode sampling mixer with a variablegain transconductance amplifier, according to a preferred embodiment ofthe present invention;

FIG. 7 is a diagram displaying a signal of interest along with severalinterferers, according to a preferred embodiment of the presentinvention;

FIG. 8 is a diagram displaying a portion of a direct RF radio receiverwith built-in close-in interference elimination through feedbackinformation, according to a preferred embodiment of the presentinvention;

FIG. 9 is a diagram displaying a schematic view of a portion of a directRF radio with built-in close-in interference elimination throughfeedback information, according erred embodiment of the presentinvention; and

FIG. 10 is a diagram displaying a signal of interest and severalinterferers, and how a band-pass filter can be used to provideinterferer prediction.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The making and using of the presently preferred embodiments arediscussed in detail below. It should be appreciated, however, that thepresent invention provides many applicable inventive concepts that canbe embodied in a wide variety of specific contexts. The specificembodiments discussed are merely illustrative of specific ways to makeand use the invention, and do not limit the scope of the invention.

The present invention will be described with respect to preferredembodiments in a specific context, namely a direct RF sampling mixeroperating in a 2.4 Gigahertz frequency band in a radio receiver that isadherent to the Bluetooth technical standards. The Bluetooth technicalstandard specifies a short-range wireless communications network whoseintended purpose is a low-power and low-cost replacement for physicalcabling. The Bluetooth technical standard is specified in a documententitled “Specification of the Bluetooth System, Version 1.1, Feb. 22,2001,” which is incorporated herein by reference. The invention may alsobe applied, however, to other wireless systems, such as globalpositioning systems (GPS), low-earth orbit satellite system basedcommunications systems, and cellular based systems that may includefirst, second, and third generation (and beyond) digital telephonesystems, time-division multiple access (TDMA), code-division multipleaccess (CDMA), global system for mobile communications (GSM) technologyalong with other digital communications technologies operating atvarious carrier frequencies. Additionally, the receiver mixer of thepresent invention has application in wired receivers as well.

With reference now to FIG. 1 a, there is shown a block diagramillustrating a prior art embodiment of a current-mode direct samplingmixer 100. The mixer 100 includes an amplifier 110 (sometimes referredto as a low-noise transconductance amplifier (LNTA), or simplytransconductance amplifier), an RF switch 115 driven by a signal 120generated by a local oscillator (not shown), and a sampling capacitor(C_(S)) 125. An alternative version of the mixer 100 exists wherein anantenna (not shown) is coupled to the amplifier, the antenna is used toreceive analog continuous-time RF signals transmitted over-the-air. Thedirect electrical coupling provides a direct signal path from theantenna into the mixer 100.

An analog continuous-time RF signal that is provided to the mixer 100(the analog RF signal may be provided to the mixer 100 via a direct wireor cable connection or transmitted over-the-air) in the form of an RFvoltage that is then converted into an RF current by the LNTA 110, whichhas a transconductance gain of g_(m). The flow of the RF current isswitched by the RF switch 115, which is driven by the signal 120generated by a local oscillator (LO). The frequency of the signal 120 isreferred to as a sampling frequency and is commonly denoted f_(S). Thesampling frequency is normally approximately equal to the frequency usedto create the analog RF signal.

As displayed in FIG. 1 a, when the signal 120 is high, the RF switch 115is closed, creating a path for the RF current. The RF current isintegrated by the sampling capacitor 125, increasing (or decreasing) thecharge on the sampling capacitor 125, depending on the direction of thecurrent flow. It is possible to view the integration of the RF currentby the sampling capacitor 125 conceptually as the injection of anelectrical charge packet that is proportional to the windowed (or gated)RF energy into the sampling capacitor 125. These electronic chargepackets increase (or decrease) the charge on the sampling capacitor 125depending on the charge polarity. In order to fully sample the analogcontinuous-time RF signal, an identical current-mode sampling mixer withan RF switch that is driven by an inverse (or complement) of the signalgenerated by the LO is used. The identical current-mode sampling mixeris used to sample the analog RF signal when the current-mode samplingmixer 100 is decoupled from the LNTA 110 by the RF switch 115 when thesignal 120 is low, as shown in FIG. 1 b.

The charge that is integrated on the sampling capacitor 125 isperiodically read out to produce a single sampled data value. Thefrequency of the charge read out can vary from being equal to thefrequency of the signal 120 to some integer divisor of the frequency ofthe signal 120. The periodic reading out of the charge on the samplingcapacitor 125 produces a discrete-time sample stream of the analog RFsignal.

Unfortunately, when the charge on the sampling capacitor 125 is beingread out, the sampling capacitor 125 cannot be used to integrate the RFcurrent, or vice versa. Therefore, the current-mode sampling mixer 100as displayed in FIG. 1 a does not permit the reading of the chargeaccumulated on its sampling capacitor 125 while the signal 120 isactively switching. Also, the amount of time required to read the chargefrom the sampling capacitor 125 is typically longer than the amount oftime to integrate the RF current, i.e., half of the period of the signal120. Therefore, it is normally not feasible to attempt a charge read outwhile the signal 120 is inactive.

Notice that the switches, both RF and non-RF switches, displayed in thefigures and discussed in this specifications are displayed as n-typemetal oxide semiconductor (NMOS) transistor switches. However, theseswitches may be made out p-type metal oxide semiconductor (PMOS) orcomplementary metal oxide semiconductor (CMOS) transistor pass gates aswell without loss in performance or generality. Of course, the use ofother types of switches may require minor rearrangements of the mixers.For example, the use of PMOS switches would require that the coupling betied to Vdd (the power source) rather than the substrate or ground asthe figures in this specifications display. However, the rearrangementsare minor and are well understood by those of ordinary skill in the artof the present invention.

With reference now to FIG. 2, there is shown a block diagramillustrating a prior art embodiment of the current-mode sampling mixer200 with cyclic charge read out. The mixer 200 is essentially the samestructurally as the mixer 150 of FIG. 1 b. When more than one samplingcapacitor is used, the current-mode sampling mixer is sometimes referredto as a multi-tap direct sampling mixer (MTDSM). A second RF switch 220and sampling capacitor 230 pair allows the task of integrating the RFcurrent to be shared between two sampling capacitors 225 and 230. The RFswitches, S1 215 and S2 220, are driven by signals 217 (for switch S1)and 222 (for switch S2). The signals 217 and 222 may be thought of asportions of the signal generated by the LO. For example, the signal 217may be configured to gate the signal produced by the LO for N cycles andthen remain low for the next N cycles and return-to gating the LO signalfor the next N cycles. The number N is equal to the number of RF cyclesthe sampling capacitors will integrate the RF current. When the twosignals 217 and 222 are combined, the result is the original signalproduced by the LO.

When one signal (217 or 222) is gating the signal produced by the LO,the RF switch (215 or 220, respectively) that is controlled by thesignal alternates between being closed and open, permitting the RFcurrent to flow to the respective sampling capacitor. When one signal(217 or 222) is gating the signal produced by the LO, the other signal(222 or 217) is low, and the switch associated with the signal is open,not permitting any RF current to reach the sampling capacitor. While onesampling capacitor is busy integrating the RF current, the secondsampling capacitor is not integrating the RF current and therefore itscharge can be read out. The roles are then reversed to allow the readingof the charge integrated by the first sampling capacitor to be read out.If the capacitance of each of the sampling capacitors is C_(S), then atany given time, the capacitance seen by the RF current remains C_(S)because the RF current only sees one sampling capacitor at a time (dueto the nature of the signals 217 and 222).

This periodic integration of a number of half-rectified RF samplesperforms a finite-impulse response (FIR) filtering operation and issometimes referred to as a temporal moving average (MA). For example, ifthe number of half-rectified RF samples being integrated in each periodis N, then the operation is referred to as a moving average N, or MA-N.The MA-N operation corresponds to an FIR filtering operation with Ncoefficients, with all coefficients being a constant value (or unity).The FIR filtering operation can be expressed in equation form as:

$w_{i} = {\sum\limits_{I = 0}^{N - 1}u_{i - l}}$Where: u_(i) is the i-th RF sample and w_(i) is the accumulated chargeon the sampling capacitor. Due to the fact that the MA-N operation isbeing read out at the lower rate of once per N RF cycles, aliasingoccurs with a foldover frequency at f₀/2N. FIR filtering and MA-Noperations are considered well understood by those of ordinary skill inthe art of the present invention and will not be discussed in detail inthese specifications.

It is important to note that the FIR filtering performed by the priorart current-mode sampling mixer 200 may be thought of as an FIR filterwith all of the filter coefficients being equal to a constant value (forexample, a one value). A constant-coefficient FIR filter is a filterwith a first zero of N-1 equidistantly spaced zeroes located at afrequency equal to f₀/N, where f₀ is the frequency of LO and N is thenumber of LO cycles that the capacitor C_(S) accumulates the RF signalprior to having the charge read-out. The constant coefficients areachieved when each of the sampling capacitors, C_(S), 225 and 230 hadthe same capacitance and the gain, g_(m), of the transconductanceamplifier 210 is held constant. It is common to desire a filter withmore zeroes in order to realize better interference elimination and/oranti-aliasing.

With reference now to FIG. 3, there is shown a data plot illustrating aseries of frequency responses for several different constant-coefficientFIR filters. A series of curves 310, 320, and 330 displays the frequencyresponses of a three different constant-coefficient FIR filters. Each ofthe three FIR filters may be the result of a single current-modesampling mixer that is operating with different parameters (such as,different filter lengths). Note the relatively large side lobes adjacentto the main lobe (the largest lobe against the left side of the dataplot). The large side lobes reduce the effectiveness of the FIR filterswhen it comes to anti-aliasing and other general filtering operations.

With reference back to FIG. 2, to achieve a FIR filter with greaterroll-off attenuation and deeper notches, the capacitance of the samplingcapacitors, C_(S), 225 and 230 can be changed as a function of timeand/or the gain, g_(m), of the transconductance amplifier 210 can varyas a function of time. For example, if either the capacitance or thegain was changed in a linear fashion, initially increasing and thendecreasing, as in a triangular saw-tooth pattern, then the FIR filteringrealized by the current-mode sampling mixer will be anarbitrary-coefficient FIR filter with double zeroes that creates doublenotching, and therefore better anti-aliasing properties near thelocations of the zeroes. Note that the manner in which the capacitanceor gain is changed can result in a different FIR filter. For example, ifthe capacitance (and gain) is maintained at a constant level, then aconstant-coefficient FIR filter is realized.

With reference now to FIG. 4 a, there is shown a diagram illustrating adirect RF sampling structure 400 wherein the capacitance of eachcapacitor seen by an analog RF signal varies with time, but the overallcapacitance at any given time remains constant, according to a preferredembodiment of the present invention. The direct RF sampling structure400, as displayed in FIG. 4 a, has five branches, although it should benoted that there is no fundamental restriction on the number of branchesin any given sampling structure and that the five branches as shown issimply a trade-off of performance and simplicity.

Each branch, for example, branch 406, is illustrated as having twocapacitors, 410 and 412 and two switches 411 and 413. Note that forbranch 406, one of the capacitors (capacitor 410) is labeled as having avalue of 0/4*C, or 0 C. Of course, this is equivalent to having nocapacitor at all. The capacitor 410 is illustrated to maintainconsistency and since in a different implementation, the capacitor 410may actually have a non-zero capacitance. The switch 411 regulates acurrent flow to the capacitor 410 while the switch 413 regulates acurrent flow to the capacitor 412. Note that for any given branch, thenet capacitance is equal. For example, branch 406 has an overallcapacitance of (0/4+4/4)*C=C. Therefore, a transconductance amplifier405 always sees a constant capacitive load. Additionally, each capacitoron each branch is grouped into one of two groups, group A and group B.For example, group A capacitors can include capacitors 410, 416, 418,420, and 422 while group B can include capacitors 412, 417, 419, 421,and 423.

The operation of the direct RF sampling structure 400 is as follows: atany given time, only one branch is coupled to the transconductanceamplifier 405, with the remaining four branches decoupled. For example,at an exemplary time, the switches 411 and 413 are closed (and allremaining switches are opened), coupling the capacitors 410 and 412 tothe transconductance amplifier 405 and accumulating a charge that isproportional to the amount of current that each capacitor receives. Notethat the branch structure is sometimes referred to as a current steeringstructure. Since the capacitors 410 and 412 have different values, thecurrent, i_(RF), is divided proportionally across the two capacitors 410and 412, depending on the capacitance of the two capacitors 410 and 412.After a period of time, the switches 411 and 413 open and another pairof switches close. This continues until all five branches have had theopportunity to accumulate current.

While the one branch is accumulating the current, one or more of theremaining branches can have its accumulated charge read off. While thecharge read-out portion of the direct RF sampling structure 400 wasomitted for simplicity's sake, it can be readily realized as a switchcoupled to each capacitor that electrically connects each capacitor to acharge read-out circuit. After the charge is read-out, any residualcharge can either be reset to zero or left on each capacitor.

A pair of switches 414 and 415 (part of a charge read-out circuit (notshown in its entirety)), controlled by control signals R_(A) and R_(B),respectively, couple the capacitors 410 and 412 to the remainder of thecharge read-out circuit. The charge read-out circuit extracts theaccumulated charge from the capacitors 410 and 412 and converts theaccumulated charge into a discrete-time sample. Note that similarswitches and charge read-out circuits exist for the remaining branches,but are not shown to maintain simplicity.

The process of current accumulation and charge read-out is continuous,meaning that once all five branches have had an opportunity toaccumulate current, the process is repeated. Note however, that asillustrated in FIG. 4, the current accumulation proceeds in aleft-to-right and then right-to-left fashion and not in a circularfashion. This is needed to prevent a large discontinuity in the value ofcapacitors, wherein there will be a sudden change in which the value ofthe capacitors will suddenly change from its maximum to its minimumvalue. As an example, the sequence of capacitors accumulating currentfrom group A could proceed as follows: capacitor 410, capacitor 416,capacitor 418, capacitor 420, capacitor 422, capacitor 420, capacitor418, capacitor 416, capacitor 410, capacitor 416, and so on. A similarsequence of capacitors for group B exists, but is not shown. Note thatthe accumulated charge by the capacitors in a branch can be read-out atany time after the charge is accumulated, but it must be completed priorto the branch being used to once again accumulate the current.

With reference now to FIG. 4 b, there is shown a diagram illustrating anexample of continuously changing capacitance seen by thetransconductance amplifier 405 (FIG. 4 a) for capacitors in group A andgroup B, wherein the change resembles a triangular saw-tooth pattern,according to a preferred embodiment of the present invention. FIG. 4 billustrates the change in the capacitance for capacitors groups A and Bif there were an infinite number of capacitors in groups A and B. Notethat when the capacitance in one group, for example, group A, increase,then the capacitance in the other group, for example, group B,decreases. Additionally, if at any given instance of time, thecapacitance of the capacitor in group A is added with the capacitor ingroup B, the result is a constant capacitance.

With reference now to FIG. 4 c, there is shown a diagram illustrating asingle period of one capacitance curve 450 for one group of capacitors,along with an overlay of a stair-step shaped curve 455 representing acapacitance curve when a finite number of capacitors (five in thisexample) is used, according to a preferred embodiment of the presentinvention. Since there can only be a finite number of capacitors that isused to reproduce the continuous capacitance curve 450, it is notpossible to exactly reproduce the continuous capacitance curve 450.However, given a sufficient number of capacitors, the approximation canbe sufficiently close. Notice that the stair-step capacitance curve 455is similar in appearance to a quantized version of the continuouscapacitance curve 450.

With reference now to FIG. 4 d, there is shown a figure illustrating theimplementation of a continuous direct RF sampling structure 459implementing an arbitrary-coefficient FIR filter using a structure of adirect RF sampling structure, according to a preferred embodiment of thepresent invention. FIG. 4 d illustrates a continuous direct RF samplingstructure 459 implementing arbitrary-coefficient FIR filtering throughthe use of transistors to steer a current as produced by atransconductance amplifier 460. Note that the direct RF samplingstructure 400 displayed in FIG. 4 a is a discrete direct RF samplingstructure, wherein switches (for example, switches 411 and 412) switchon and off to regulate current flow. The continuous direct RF samplingstructure 459 is continuous in that the current is continually flowingthrough the transistors when a switch 465 is closed and permits thecurrent to flow.

A pair of transistors 472 and 474 operates as a current steeringdifferential pair. Each of the two transistors 472 and 474 may becontrolled by differentially opposed voltages, so that the transistors472 and 474 can regulate the flow of a current produced by atransconductance amplifier 460. Depending on the state of the twotransistors 472 and 474, a varying amount of current can flow throughboth branches to capacitors 475 and 480 is under the constraint thattheir sum is equal to i_(RF) (the current produced by thetransconductance amplifier 460). Examples of the voltages that may beused to drive the transistors 472 and 474 are displayed as voltages 490and 495.

According to the magnitudes of the voltages 490 and 495 at thetransistors 472 and 474, the current is split into proportional partsaccording to the saw-tooth transfer function (as displayed in voltages490 and 495). For example, if one transistor passes 10% of the current,i_(RF), then the other transistor will pass the remaining 90% of thecurrent, but the total current passed by the two transistors 472 and 474combined would be the same. The capacitors 475 and 480 would thenaccumulate a charge that is proportional to the amount of current thateach capacitor receives. Note that in order to present constantcapacitative load to the RF source, both capacitors should have the samecapacitance. According to a preferred embodiment of the presentinvention, it is also possible to duplicate the switch 465 and thenrelocate the switches (not shown) to a position below the switches 472and 474 and so that one switch is directly coupled to each capacitor 475and 480. The two switches would remain controlled by the signalgenerated by the LO. The resulting direct RF sampling structure wouldoperate in a similar to the direct RF sampling structure describedabove.

A pair of switches 481 and 482 (part of a charge read-out circuit (notshown in its entirety)), controlled by control signals R_(A) and R_(B),respectively, couple the capacitors 475 and 480 to the remainder of thecharge read-out circuit. The charge read-out circuit extracts theaccumulated charge from the capacitors 475 and 480 and converts theaccumulated charge into a discrete-time sample.

The number of branches in a direct RF sampling structure can beincreased to achieve a better approximation of the continuouscapacitance curves. However, as the number of branches exceeds a certainnumber, perhaps six or seven, it becomes unwieldy to create a largenumber of branches. Additionally, producing a large number of capacitorswith each capacitor having essentially a different capacitance can bedifficult. A more regular pattern is needed if an extra-ordinarily largenumber of capacitors are to be used in the direct RF sampling structure.

With reference now to FIG. 5, there is shown a direct RF samplingstructure 500 that can support an extra-ordinarily large number ofcapacitors to provide an excellent approximation of a continuouscapacitance curve, according to a preferred embodiment of the presentinvention. Rather than arranging pairs of capacitors and switches inbranches, the capacitors and switches are arranged in an array 520 ofcapacitors and switches. Note that each capacitor in the array 520 hasthe same capacitance, making it much easier to create the capacitors andthe array 520. The array 520 is made up of columns of j capacitors (forexample, column 521) and rows of i capacitors (for example, row 522)arranged in a cross-point grid. The array 520, therefore, has a total ofi*j capacitors and switches.

Highlight 510 provides a detailed view of a single capacitor 515 andswitch 517 combination. The capacitor 515 is coupled to the cross-pointgrid via the switch 517. The switch 517 is, in turn, controlled by acontrol signal (not shown). Each capacitor can be addressed by itscolumn and row index, for example, capacitor 515 may be addressed as(6,6), assuming that a capacitor with an address (0,0) is at the upperleft hand corner of the array 520.

The operation of the direct RF sampling structure 500 is similar to thedirect RF sampling structure 400 (FIG. 4 a), with only one column (forexample, column 521) being coupled to a transconductance amplifier 505at any given time, and the remaining columns being decoupled. However,in the direct RF sampling structure 500, everyone of the j capacitors inthe column accumulate the current. Once one column has accumulated thecurrent for a specified amount of time, the column becomes decoupledfrom the transconductance amplifier 505 and a column adjacent to thatcolumn becomes coupled to the transconductance amplifier 505. Theprocess continues, with the coupled column sweeping across the array520.

After a column accumulates the current for the specified amount of time,the charge accumulated is read-out. For any given column, a specifiednumber of capacitors is coupled to an A buffer 530 and the remainingcapacitors in the row are coupled to a B buffer 535. For example, incolumn 521, one capacitor may be coupled to the A buffer 530 while theremaining six capacitors are coupled to the B buffer 535. Therefore, theA buffer 530 receives a charge proportional to 1/7^(th) of the totalcharge accumulated, while the B buffer 535 receives a chargeproportional to 6/7^(th) of the total accumulated charge. In the columnadjacent to row 521, the A buffer 530 may receive 2/7^(th) of the charge(two capacitors) and the B buffer 535 receives 5/7^(th) of the totalcharge (five capacitors), and so on.

According to a preferred embodiment of the present invention, the numberof capacitors coupled to a particular buffer (530 or 535) can beconfigured dynamically, so that the FIR filter coefficients can bechanged on the fly. According to yet another preferred embodiment of thepresent invention, the array 520 can be replaced with a charge coupleddevice (CCD) (not shown) of similar topology.

Charge coupled devices (CCDs) are metal oxide semiconductor (MOS)capacitors formed into a linear string. CCDs can store and transferanalog charge signals and can be used for various types of signalprocessing applications such as electronically variable delay lines andtransversal filters. CCDs store and transfer charge (which representsinformation) between potential wells at or near the surface of thesubstrate. A charge is placed in the potential wells by applying avoltage across the wells. This charge can represent information carriedon the current provided by the transconductance amplifier. CCDs are wellknown by those of ordinary skill in the art of the present invention.

An additional method that can be used to effectively change the FIRcoefficients is to vary the gain, g_(m), of the transconductanceamplifier. When the gain of the transconductance amplifier is varied ina desired pattern and the capacitance seen by the transconductanceamplifier is kept constant, the net effect is that a varying amount ofcharge is accumulated onto the capacitors, hence yielding arbitrary FIRcoefficients.

With reference now to FIG. 6, there is shown a schematic of a direct RFsampling structure 600 wherein a gain of a transconductance amplifier610 is varied according to a specified pattern to obtainarbitrary-coefficient FIR filtering, according to a preferred embodimentof the present invention. The direct RF sampling structure 600 issimilar to the direct RF sampling structure 200 (FIG. 2) with theexception of a variable gain transconductance amplifier 610. A gaincurve 612 displays an exemplary trace of the gain of thetransconductance amplifier 610. A pair of switches 615 and 620, drivenby signals S1 617 and S2 622, gate the output of the transconductanceamplifier 610 to a pair of capacitors 625 and 630. According to apreferred embodiment of the present invention, the capacitors 625 and630 have equal capacitance. The charge accumulated on the capacitors 625and 630 can be read-out by charge read-out circuits (not shown) when thecapacitor is not actively accumulating the current.

Note that the gain of the transconductance amplifier 610 must be set sothat at its maximum, there remains sufficient dynamic range so that aninput signal does not cause the transconductance amplifier 610 to clip.Clipping of the input signal would result in loss of information anddistortion, with the amount of information depending on the degree ofclipping.

The removal of out-of-band interferers can be achieved through the useof analog filters. These analog filters can be located as early in aradio receiver as immediately after an antenna. If the out-of-bandinterferers are located a relatively large distance (in frequency terms)away from a signal of interest, then it is possible to use low-orderanalog filters that are relatively easy to implement out of a relativelysmall number of components. However, if the out-of-band interferers areclose to the signal of interest (commonly referred to as close-ininterferers), then high-order analog filters are required to eliminatethe interferers. High-order analog filters can be hard to implement inan integrated circuit and at the very least, they can consume asignificant amount of real-estate.

If the close-in interferers cannot be eliminated, then it is common torequire that RF front-end electronics have a good degree of linearity tohandle the potentially large dynamic range of the both the signal ofinterest and the interferers. For example, it is common to encounterinterferers that are several orders of magnitude larger than the signalof interest. Therefore, in order to prevent loss of importantinformation in the signal of interest, RF front-end electronics musthave sufficient linearity to process both the signal of interest and theinterferers without (significant) distortion. This requirement forlinearity raises the overall cost of the radio receiver.

With reference now to FIG. 7, there is shown a diagram illustrating asignal of interest 705 along with some interferers, according to apreferred embodiment of the present invention. The signal of interest705 is shown adjacent to two close-in interferers 710 and 715 and afar-away interferer 720. Note that as illustrated, the magnitudes ofsome of the interferers, namely interferers 710 and 720 aresignificantly greater than the magnitude of the signal of interest 705.As stated above, the interferers may be several orders of magnitude (ormore) larger than the signal of interest 705. However, FIG. 7 does notdisplay this behavior.

Also shown in FIG. 7 is a frequency response of a low-order low-passfilter (LPF) 735 that can be used to eliminate the far away interferer720. Additionally, a frequency response of a high-order LPF 730 that canbe used to eliminate all three of the interferers. Note that since theclose-in interferer 710 is so close to the signal of interest 705, itmay not be possible to completely eliminate the close-in interferer 710with the high-order LPF 730 as displayed. Perhaps it would be possibleto use a higher order LPF (not shown) to completely eliminate theclose-in interferers.

As an alternative to using analog filters in a radio receiver's analogfront-end to eliminate the close-in interferers (the far awayinterferers can be easily eliminated by a low-order LPF), an iterativetechnique using feedback information and digital filters can be used toremove the close-in interferers and linearize a sampling structure of adirect RF radio receiver. By linearizing the sampling structure, thelinearity of the analog front-end can be relaxed, resulting in loweroverall power consumption and a less expensive direct RF radio receiver.A basic idea of the technique is to use digital filters in place ofanalog filters, where it is easy and inexpensive to create high-orderfilters.

With reference now to FIG. 8, there is shown a block diagram of aportion of a direct RF radio receiver 800 with built-in close-ininterference elimination through feedback information, according to apreferred embodiment of the present invention. For a more detailedexplanation of the operation of the basic direct RF radio receiver,refer to a related co-pending and co-assigned patent application Ser.No. 10/190,867, filed Jul. 08, 2002, entitled “Direct Radio Frequency(RF) Sampling with Recursive Filtering Method”, which is incorporatedherein by reference.

Briefly, the direct RF radio receiver 800 has as its input an analogcontinuous-time RF signal. The analog RF signal may or may not have beenfiltered to eliminate some of the out-of-band signals. A sampling unit805 preferably uses a current-mode sampling mixer to create adiscrete-time sample stream from the analog RF signal. The discrete-timesample stream is then filtered, buffered, and converted into a digitalbitstream by a signal processing unit 815. The digital bitstream is thenprovided to digital circuitry that provides further processing of thesignal to turn it into a usable form for a digital processor (not shown)attached to the direct RF radio receiver 800. The sampling unit 805 iscontrolled by a digital control unit (DCU) 810 that is responsible forgenerating control and timing signals.

The built-in close-in interference elimination is provided via aninterference predictor 840, whose function is to predict the interferersand provide the information regarding the interferers to a feedbackcircuit 845. The feedback circuit 845 uses the information from theinterference predictor 840 and using destructive combination, eliminatesthe interferers in the sampling unit 805. Through an iterative processof predicting the interferers in the interferer predictor 840 andeliminating them in the sampling unit 805, the technique ends uplinearizing the sampling unit 805.

For example, a first time through the direct RF radio receiver 800, whenthere is no feedback information about any interferers, all interfererspass through to an output of the signal processing unit 815. It is onlyafter the first time that the digitized version of the analog RF signalgoes through the direct RF radio receiver 800 that the interfererpredictor 840 is able to produce any information about interferers thatmay exist in the digital bitstream. This information is then passed backto the sampling unit where it is eliminated from the discrete-timesample stream.

According to a preferred embodiment of the present invention, theinterferer predictor 840 can be as simple as a band-pass filter (BPF) orhigh-pass filter (HPF) that is set to reject the signal of interest andfilter out the interferers. The BPF or the HPF can be digital filters ofa needed order to ensure that the signal of interest is completelyeliminated. Since digital filters are “software” filters (not resistors,capacitors, and inductors), it is relatively easy to create digitalfilters of arbitrary order. By choosing to eliminate the signal ofinterest, the interferer predictor 740 may provide feedback informationabout every component other than the signal of interest existing in thedigital bitstream. Alternatively, the interferer predictor 840 may be apredictor in general and specifically, a linear predictor. Predictors,linear predictors and the use of BPFs and HPFs to produce informationabout interferers are well known by those of ordinary skill in the artof the present invention.

Due to the fact that the sampling unit 805 is a current-mode samplingmixer, simple mathematical subtraction of the feedback information doesnot apply to the elimination of any predicted interferers. Rather, thefeedback information must be accumulated by sampling capacitors in thesampling unit 805. It is through charge sharing that the feedbackinformation is used. Please refer to a co-pending and co-assigned patentapplication Ser. No. 10/147,784, filed May 16, 2002, entitled “EfficientCharge Transfer Using a Switched Capacitor Resistor” for a more detailedexplanation of how the feedback circuitry makes use of feedbackinformation.

The feedback information provided to the sampling unit 805 for close-ininterferer rejection can be over-sampled and noise shaped. In thisconfiguration, a DAC (which can be a part of the interferer predictor840 or the feedback circuit 845) can be a sigma-delta digital-to-analogconverter which provides noise shaped output to the sampling unit 805.The filtering in the receiver is used to reject the shaped quantizationnoise as it is already there.

With reference now to FIG. 9, there is shown a schematic diagram of aportion of the direct RF radio receiver 800 (FIG. 8), with detailedviews of the sampling unit 805, DCU 810, and feedback circuit 845,according to a preferred embodiment of the present invention. Note thatthe schematic diagram displays only a portion of a complete direct RFradio receiver 800. A buffer 950 represents an output buffer of thesampling unit 805. What is shown is an LO+ and LO− portion of either anin-phase (I) or quadrature-phase (Q) signal path for a direct RF radioreceiver operating in differential mode. The LO+ and the LO− (for eitherthe I or Q signal paths) portions are essentially similar and adescription of one portion will also sufficiently describe the other.

Once again, detailed explanations of the operation of the sampling unit805 and the DCU 810 can be found in the related patent application Ser.No. 10/190,867, filed Jul. 08, 2002, entitled “Direct Radio Frequency(RF) Sampling with Recursive Filtering Method”.

Taking a closer look at the feedback circuit 845, there is a currentsource 905 that is used to represent the feedback information, in theform of a feedback current, i_(FBCK), provided by the interfererpredictor 840 (FIG. 8, not shown in FIG. 9). The feedback current isaccumulated by a pair of capacitors, C_(F), 912 and 913, wheneverswitches 910 and 911 are closed. The switches are controlled by signalsgenerated by the DCU 810. One of the capacitors is used to accumulatethe feedback current for one capacitor bank 920 of the sampling unit 805and the other capacitor is used to accumulate the feedback current forthe other capacitor bank 921 of the sampling unit 805. The two feedbackcapacitors 912 and 913 are coupled to the sampling unit 805 by switches914 and 915.

After one the two capacitors 912 or 913 has accumulated the feedbackcurrent for a specified amount of time, one of the two switches 914 or915 is closed and the capacitor is coupled to the correspondingcapacitor bank of the sampling unit 805, and the charge accumulated bythe capacitor is shared with the capacitors in the capacitor bank. Thisprocess operates continuously to provide the feedback information toboth capacitor banks of the sampling unit 805.

With reference now to FIG. 10, there is shown a diagram illustrating asignal of interest 1005 along with several interferers 1010, 1015, and1020, and how a band-pass filter can be used to provide interfererinformation, according to a preferred embodiment of the presentinvention. The signal of interest 1005 is shown with the pair ofclose-in interferers 1010 and 1015 and the far away interferer 1020.Also shown is a frequency response 1025 of a band-pass filter thateliminates everything outside of its lines. Therefore, a majority of thesignal of interest 905 is eliminated and the far-away interferer 1020 isalso eliminated. The band-pass filter retains the two close-ininterferers 1010 and 1015. A signal conveying this information isfedback to the sampling unit via charge accumulation and then chargesharing so that the two close-in interferers 1010 and 101 5 can beeliminated from the discrete-time sample stream that is generated by thesampling unit.

Although the present invention and its advantages have been described indetail, it should be understood that various changes, substitutions andalterations can be made herein without departing from the spirit andscope of the invention as defined by the appended claims.

Moreover, the scope of the present application is not intended to belimited to the particular embodiments of the process, machine,manufacture, composition of matter, means, methods and steps describedin the specification. As one of ordinary skill in the art will readilyappreciate from the disclosure of the present invention, processes,machines, manufacture, compositions of matter, means, methods, or steps,presently existing or later to be developed, that perform substantiallythe same function or achieve substantially the same result as thecorresponding embodiments described herein may be utilized according tothe present invention. Accordingly, the appended claims are intended toinclude within their scope such processes, machines, manufacture,compositions of matter, means, methods, or steps.

1. A method for providing filtering in a current-mode sampling mixercomprising: 1) providing a received radio frequency (RF) current; 2)coupling the RF current to a first load for a first period of time toaccumulate a charge on the first load; 3) decoupling the RF current fromthe first load; 4) coupling the RF current to a second load for a secondperiod of time to accumulate a charge on the second load, wherein thefirst and second loads are capacitive loads with substantially equalcapacitance, wherein each load comprises at least two dynamicallyswitchable capacitors, and the capacitors may have unequal capacitances;5) reading the accumulated charge on the first load; 6) decoupling theRF current from the second load; 7) reading the accumulated charge onthe second load; and 8) repeating 2–8.
 2. The method of claim 1, whereinthe RF current is output by an amplifier and the amplifier has variablegain that varies with time.
 3. The method of claim 2, wherein the gainof the amplifier varies in a periodic and continuous manner.
 4. Themethod of claim 3, wherein at a maximum, the gain of the amplifier hassufficient dynamic range to ensure that the RF current does not clip. 5.The method of claim 1, wherein the loads are reset after having theiraccumulated charges read-out.
 6. A method for providing filtering in acurrent-mode sampling mixer comprising: 1) providing a received radiofrequency (RF) current; 2) coupling the RF current to a first load for afirst period of time to accumulate a charge on the first load; 3)decoupling the RF current from the first load; 4) coupling the RFcurrent to a second load for a second period of time to accumulate acharge on the second load; 5) reading the accumulated charge on thefirst load; 6) decoupling the RF current from the second load; 7)reading the accumulated charge on the second load; and 8) repeating 2–8,wherein there are two capacitors per load, wherein a first capacitor ofeach load when sequentially ordered in a coupling order to the RFcurrent, the capacitances increase from left to right, and wherein asecond capacitor of each load when sequentially ordered in the couplingorder to the RF current, the capacitances decrease from left to right.7. The method of claim 6, wherein the capacitances of the firstcapacitors of each load trace out a periodic waveform withtriangular-shaped periods.
 8. The method of claim 6, wherein the RFcurrent is output by an amplifier and the amplifier has variable gainthat varies with time.
 9. The method of claim 8, wherein the gain of theamplifier varies in a periodic and continuous manner.
 10. The method ofclaim 9, wherein at a maximum, the gain of the amplifier has sufficientdynamic range to ensure that the RF current does not clip.
 11. Themethod of claim 6, wherein the loads are reset after having theiraccumulated charges read-out.
 12. A method for providing filtering in acurrent-mode sampling mixer comprising: 1) providing a received radiofrequency (RF) current; 2) coupling the RF current to a first load for afirst period of time to accumulate a charge on the first load; 3)decoupling the RF current from the first load; 4) coupling the RFcurrent to a second load for a second period of time to accumulate acharge on the second load; 5) reading the accumulated charge on thefirst load; 6) decoupling the RF current from the second load; 7)reading the accumulated charge on the second load; and 8) repeating 2–8,wherein there are a total of L loads, numbered 0, 1, to L-1, wherein theloads are coupled to the RF current and read in a specified order, andwherein the specified order is: 0, 1, . . . , L-2, L-1, L-2 , . . . , 1,0.
 13. The method of claim 12, wherein the RF current is output by anamplifier and the amplifier has variable gain that varies with time. 14.The method of claim 13, wherein the gain of the amplifier varies in aperiodic and continuous manner.
 15. The method of claim 14, wherein at amaximum, the gain of the amplifier has sufficient dynamic range toensure that the RF current does not clip.
 16. The method of claim 12,wherein the loads are reset after having their accumulated chargesread-out.
 17. A method for providing filtering in a current-modesampling mixer comprising: 1) providing a received radio frequency (RF)current; 2) coupling the RF current to a first load for a first periodof time to accumulate a charge on the first load; 3) decoupling the RFcurrent from the first load; 4) coupling the RF current to a second loadfor a second period of time to accumulate a charge on the second loadwherein the first and second loads are capacitive loads withsubstantially equal capacitance, wherein each load comprises at least Jdynamically switchable capacitors, and wherein all capacitors have equalcapacitances; 5) reading the accumulated charge on the first load; 6)decoupling the RF current from the second load; 7) reading theaccumulated charge on the second load; and 8) repeating 2–8.
 18. Themethod of claim 17, wherein in the first and second couplings, all Jcapacitors accumulate a charge based on the RF current.
 19. The methodof claim 17, wherein the RF current is output by an amplifier and theamplifier has variable gain that varies with time.
 20. The method ofclaim 19, wherein the gain of the amplifier varies in a periodic andcontinuous manner.
 21. The method of claim 20, wherein at a maximum, thegain of the amplifier has sufficient dynamic range to ensure that the RFcurrent does not clip.
 22. The method of claim 17, wherein the loads arereset after having their accumulated charges read-out.
 23. A method forproviding filtering in a current-mode sampling mixer comprising: 1)providing a received radio frequency (RF) current; 2) coupling the RFcurrent to a first load for a first period of time to accumulate acharge on the first load; 3) decoupling the RF current from the firstload; 4) coupling the RF current to a second load for a second period oftime to accumulate a charge on the second load; 5) reading theaccumulated charge on the first load wherein W capacitors are coupled toa first charge read-out circuit and X capacitors are coupled to a secondcharge read-out circuit; 6) decoupling the RF current from the secondload; 7) reading the accumulated charge on the second load wherein Ycapacitors are coupled to the first charge read-out circuit and Zcapacitors are coupled to a second charge read-out circuit, whereinW+X=J and Y+Z=J, and wherein the first and second loads are capacitiveloads with equal capacitance, wherein each load comprises at least Jcapacitors which accumulate a charge based on the RF current, andwherein all capacitors have equal capacitances; and 8) repeating 2–8.24. The method of claim 23, wherein the loads are charge-coupled devices(CCDs).
 25. The method of claim 23, wherein the RF current is output byan amplifier and the amplifier has variable gain that varies with time.26. The method of claim 25, wherein the gain of the amplifier varies ina periodic and continuous manner.
 27. The method of claim 26, wherein ata maximum, the gain of the amplifier has sufficient dynamic range toensure that the RF current does not clip.
 28. The method of claim 23,wherein the loads are reset after having their accumulated chargesread-out.
 29. A current-mode sampling mixer comprising: an amplifier toproduce a current based on a radio frequency (RF) signal; a plurality ofloads, each load switch-ably coupled to the amplifier, each loadcontaining circuitry to accumulate a charge based on the current whereineach load comprises at least two dynamically switchable capacitors whichmay be of unequal capacitance; and a plurality of charge read-outcircuits, each charge read-out circuit switch-ably coupled to a load,the charge read-out circuit to permit extraction of the chargeaccumulated on the load.
 30. The current-mode sampling mixer of claim29, wherein the switch is controlled via a control signal, and whereinwith the switch closed, the capacitor receives a portion of the currentfrom the amplifier based on its capacitance.
 31. The current-modesampling mixer of claim 29, wherein the capacitors in each load aresized so that an overall capacitance of each load is equal.
 32. Thecurrent-mode sampling mixer of claim 29, wherein each load comprises atleast J capacitors of equal capacitance.
 33. The current-mode samplingmixer of claim 32, wherein the loads are charge-coupled devices (CCDs).34. A current-mode sampling mixer comprising: an amplifier to produce acurrent based on a radio frequency (RF) signal; a plurality of loads,each load switch-ably coupled to the amplifier and including twocapacitors to accumulate a charge based on the current and wherein afirst capacitor of each load has a capacitance so that when sequentiallyarranged in order of coupling to the amplifier, the capacitancesincrease from left to right, and wherein a second capacitor of each loadhas a capacitance so that when sequentially arranged in order ofcoupling to the amplifier, the capacitances decrease from left to right;and a plurality of charge read-out circuits, each charge read-outcircuit switch-ably coupled to a load, the charge read-out circuit topermit extraction of the charge accumulated on the load.
 35. Thecurrent-mode sampling mixer of claim 34, wherein there are L totalloads, numbered from 0, 1, to L-1, and wherein the coupling order to theamplifier is specified as: 0, 1, . . . , L-2, L-1, L-2 , . . . , 1, 0.36. The current-mode sampling mixer of claim 34, wherein the couplingorder to the amplifier is repeated.
 37. A current-mode sampling mixercomprising: an amplifier to produce a current based on a radio frequency(RF) signal; a plurality of loads, each load switch-ably coupled to theamplifier, wherein there are two charge read-out circuits per load,wherein for each load, at least J capacitors of equal capacitance toaccumulate a charge based on the current coupled to a first chargeread-out circuit plus at least J capacitors of equal capacitance toaccumulate a charge based on the current coupled to a second chargeread-out circuit is equal to J, and wherein the number of capacitorscoupled to the first and second charge read-out circuits can bedifferent per different load; and a plurality of charge read-outcircuits, each charge read-out circuit switch-ably coupled to a load,the charge read-out circuit to permit extraction of the chargeaccumulated on the load.
 38. The current-mode sampling mixer of claim37, wherein when arranged in order of coupling to the amplifier, thenumber of capacitors coupled to the first charge read-out circuitincreases from left to right, and wherein when arranged in order ofcoupling to the amplifier, the number of capacitors coupled to thesecond charge read-out circuit decreases from left to right.
 39. Acurrent-mode sampling mixer comprising: a variable gain amplifier thatvaries with time for producing a current based on a radio frequency (RF)signal; a plurality of loads, each load switch-ably coupled to theamplifier, each load containing circuitry to accumulate a charge basedon the current; and a plurality of charge read-out circuits, each chargeread-out circuit switch-ably coupled to a load, the charge read-outcircuit to permit extraction of the charge accumulated on the load. 40.The current-mode sampling mixer of claim 39, wherein each load is asingle capacitor, and wherein each load has an equivalent capacitance.41. The current-mode sampling mixer of claim 39, wherein the gain of theamplifier varies in a periodic and continuous manner.